SEMICONDUCTOR VOLTAGE REGULATOR FOR AN ALTERNATING-CURRENT GENERATOR

B. V. Lukutin, A. B. Tsukublin

(Presented by the scientific seminar of the Departments of Electrical Machines and General Electrical Engineering)

In connection with the expansion of the field of application of thyristor converters and the creation on their basis of machine–converter complexes, the problem arises of creating fast-acting voltage regulators operating with a nonsinusoidal form of the generator-voltage curve. For alternating-current generators the fundamental question is by what value of the voltage regulation should be carried out: by the amplitude, the mean, or the effective value. Depending on the solution of this question, the circuit of the sensing element is selected. It should be noted that, with a sinusoidal curve form, sufficient stability of the alternating voltage can be achieved by a regulation system for any of its values [1, 2].

In practice it is often necessary to obtain a stable effective value of the nonsinusoidal voltage of a synchronous generator over a wide range of load and frequency variation. In this case the sensing element must respond to the effective value of the nonsinusoidal alternating voltage:

\[ U=\sqrt{\frac{1}{T}\int_{0}^{T}u^{2}\,dt}. \]

It follows from this expression that the effective-value sensor must be a computing device that successively performs the operations of squaring the instantaneous value of the voltage, integrating, and extracting the square root. That is, the effective-value sensor turns out to be rather complex, unreliable, and not providing high measurement accuracy. Of considerable interest in this connection are methods for measuring the effective value that use the effect of thermal action. Of these, the simplest and most accurate is the use, as an effective-value sensor, of an incandescent lamp, which converts the generator voltage into a luminous flux that is registered by a photoreceiver. The photoreceiver may be photoresistors, photodiodes, phototransistors, etc. The incandescent lamp is a universal controllable light source for a wide class of optoelectronic pairs. It makes it possible to easily obtain high illumination levels on the working area of the photoreceiver, which reduces the requirements on its sensitivity and gain factor [3].

In the present article a voltage regulator for an alternating-current generator is described, in which, as the effective-value sensor,

an optoelectronic pair is used: an incandescent lamp–photoresistor, connected in a bridge circuit (Fig. 1a). The voltage \(U_c\), proportional to the generator voltage, is taken from one diagonal of the bridge (Fig. 1a), while the supply voltage \(U_0\) is applied to the other diagonal. A photoresistor \(r_\phi\), which operates

Fig. 1. RMS-value sensor: a — circuit;
b — dependence of the photoresistor resistance on the voltage across the incandescent lamp

in the controlled-resistance mode, is included in one arm of the bridge. The dependence of the photoresistor resistance on the voltage across the incandescent lamp is shown in Fig. 1b. The generator voltage, and correspondingly the voltage across the incandescent lamp, varies within a narrow range from \(U_{\min}\) to \(U_{\max}\); therefore the working portion of the dependence \(r_\phi=f(U_\ell)\) may be approximated by a straight line with the corresponding slope coefficient. Then, in accordance with Fig. 1,

\[ U_c=\frac{U_0 r_\phi}{r_3+r_\phi}-\frac{U_0 r_1}{r_1+r_2}, \]

where

\[ r_\phi=r' + (U_{\max}-U_\ell)\operatorname{tg}\alpha; \]

\[ \operatorname{tg}\alpha=\frac{r''-r'}{U_{\max}-U_{\min}}. \]

Consequently, the change in the bridge voltage \(\Delta U_c\) due to a change in the lamp voltage \(\Delta U_\ell\) is determined by the expression

\[ \Delta U_c= \frac{U_\ell\left(r_{\mathrm{n}}+\Delta U_\ell \operatorname{tg}\alpha\right)} {r_3+r_{\mathrm{n}}+\Delta U_\ell \operatorname{tg}\alpha} - \frac{U_0 r_1}{r_1+r_2}. \tag{1} \]

Expression (1) shows that the magnitude \(\Delta U_c\) can be significant, and that the ratio of the voltages \(U_c\) of the unbalanced and balanced bridge can attain very large values. Consequently, the sensitivity of the regulator can be sufficiently high.

The schematic diagram of the regulator is shown in Fig. 2. The voltage from the bridge diagonal is applied to the input of an emitter follower assembled on transistors \(T_1, T_2\). To increase the input resistance of the follower, a Darlington circuit is used. From the output of the emitter follower, the unbalance voltage is fed to the base circuit of transistor \(T_3\). Positive sawtooth pulses from a sawtooth-voltage generator (SVG), assembled on transistor \(T_8\), are also fed here. The SVG is triggered by a multivibrator assembled on transistors \(T_6, T_7\). These voltages are compared across resistor \(r_6\). Transistor \(T_3\) is cut off as soon as the sawtooth voltage

Fig. 2. Schematic circuit diagram of the voltage regulator

will become equal to the bridge unbalance voltage and opens as soon as the unbalance voltage exceeds the sawtooth voltage. Thus, transistor \(T_3\) operates as a typical relay element, converting the smoothly varying signal arriving from the emitter follower into a series of pulses whose duty factor \(K_{\mathrm{d}}\) is proportional to the magnitude of this signal. The duty factor is determined by the expression:

\[ K_{\mathrm{d}}=\frac{t_{\mathrm{p}}}{T}, \]

where \(t_{\mathrm{p}}\) is the time during which the current pulse passes in the collector of transistor \(T_3\);

\(T\) is the period of the operating frequency of the sawtooth-voltage generator.

The relation between the change in the magnitude of the bridge unbalance signal and the corresponding change in pulse duration is shown in Fig. 3. Let us consider the case in which, as a result of a change in the generator load, its output voltage has changed; this means that the bridge unbalance voltage has also changed from \(U_{c1}\) to \(U_{c2}\). Consequently, the magnitude of the duty factor \(K_{\mathrm{d}}\) will also change. The change in pulse duration when the generator voltage changes is determined as

Fig. 3

\[ \Delta t_i = t_{i2} - t_{i1}, \]

where \(t_{i1}\) is the pulse duration corresponding to the initial generator voltage;

or

\[ \Delta t_i = \Delta U_c \left( \frac{1}{\operatorname{tg}\beta} + \frac{1}{\operatorname{tg}\gamma} \right), \]

where

\[ \operatorname{tg}\beta = \frac{U_m}{t_p}, \]

\[ \operatorname{tg}\gamma = \frac{U_m}{t_v}; \]

\(U_m\) is the amplitude of the sawtooth-voltage pulses;
\(t_p\) is the working stroke of the sawtooth;
\(t_v\) is the recovery period of the initial state of the sawtooth-voltage generator.

Usually \(t_p \gg t_v\),

\[ \Delta t_i \simeq \frac{\Delta U_c}{\operatorname{tg}\beta}. \tag{2} \]

Therefore, the duration of the current pulse in the collector load depends linearly on the magnitude of the signal increment \(\Delta U_c\), and then the new value of \(K_{zi}\) is determined as

\[ K_{zi} = \frac{1}{T}\left( \frac{\Delta U_c}{\operatorname{tg}\beta} + t_{i1} \right), \]

where \(\Delta U_c\) is determined by expression (1).

Thus, the sensitivity of the regulator is determined by the magnitude of \(\Delta U_c\) and by the angle \(\beta\), and to reduce the error it is necessary to increase \(\Delta U_c\), which is entirely feasible for a bridge circuit, and to reduce the amplitude of the sawtooth \(U_m\).

The duty factor corresponding to the nominal value of the regulated voltage is set by means of potentiometer \(r_2\). The positive-feedback circuit \(c_1\)--\(r_7\) provides sharp fronts during switching of the amplifier. Bias \(E_3\) ensures reliable and rapid turnoff of the power transistor \(T_5\). From the output of the power amplifier, the pulses are applied to the field winding of the electric machine, where they are demodulated by means of the shunting diode \(D_2\). The current in the field winding, with a sufficiently large time constant, has practically no ripple, and its magnitude is determined by the expression

\[ I_{\mathrm{ovg}} = \frac{E_2}{r_{\mathrm{ovg}}}\cdot K_{zi}, \]

where \(r_{\mathrm{ovg}}\) is the resistance of the generator field winding.

Diode \(D_1\) serves to remove overvoltages from transistor \(T_5\), and resistor \(r_{12}\) shunts it, facilitating the operating mode of the power transistor.

A regulator prototype built according to the circuit described provided an accuracy of regulation of the generator voltage of \(\pm 2\%\) when the generator load was varied from zero to nominal and when the generator rotation speed was varied twofold. A semiconductor rectifier was used as the generator load.

For the optocoupler pair consisting of an incandescent lamp and a photoresistor, SF-2-5 photoresistors and incandescent lamps were used.

of the SM-36 type. The resistance of the photoresistor in the operating section was 20–30 kΩ.

The present article has described a voltage regulator that provides stabilization of the rms value of the nonsinusoidal voltage at the generator output with an accuracy of ±2% over a wide range of changes in the load and in the generator rotation speed. The regulator is distinguished by sufficient simplicity, reliability, and good weight-and-size characteristics.

REFERENCES

1. V. P. Bunakov, R. G. Gasparov. Semiconductor voltage and frequency regulators for electric machines. Energiya Publishing House, 1965.

2. E. A. Lodochnikov, E. G. Orlov. On the calculation of a pulse-control system for electric machines with an implicitly expressed pulse-width modulator. Elektrichestvo, 1972, No. 8.

3. S. V. Svechnikov. Elements of optoelectronics. Moscow, Energiya Publishing House, 1971.